This invention relates to a transmission power control apparatus of a base station for controlling the transmission power of a mobile station in a CDMA mobile communication system. More particularly, the invention relates to a transmission power control apparatus for obtaining a desired value of BER (Bit Error Rate) by correcting a SIR (target SIR), which is the target of transmission power control, based upon fading pitch or level differences between paths.
Analog schemes were used as modulation schemes for mobile communication in the past but present-day schemes are mainly digital. In general, an analog cellular scheme is referred to as a first-generation scheme, and a digital scheme such as PDC (the Japanese standard), GSM (the European standard), IS54 (the TDMA standard in the USA) and IS95 (the CDMA standard in the USA) is referred to as a second-generation scheme. Voice service is the focus up to the second generation, in which full use is made of analog/digital narrow-band modulation/demodulation to achieve communication by making effective utilization of the limited radio band.
In next-generation schemes, however, it will be possible to perform not only telephone conversation but also communication by facsimile and electronic mail, etc., and communication between computers. In order to achieve this, a desired communication scheme for the next generation will be one in which various information (multimedia information) services for moving and still images will be possible in addition to voice and information provided by communication means, and in which high-quality communication is made possible in such a manner that the mobile network is transparent to the user. DS-CDMA (Direct Sequence Code Division Multiple Access) communication is the focus of attention as a promising candidate for next-generation wireless access. Such a DS-CDMA communication scheme achieves spectrum spreading by directly multiplying a signal, which is to undergo spectrum spreading, by a signal having a band much broader than that of the first-mentioned signal.
FIG. 24 is a block diagram showing a CDMA receiver having a diversity construction in which outputs from respective ones of branches are combined by maximal ratio combining and data is discriminated based upon the combined results. Each of branches B1 and B2 has a radio unit 11 for converting a high-frequency signal received by an antenna 10 to a baseband signal by applying a frequency conversion (RF→IF conversion). A quadrature detector 12 subjects the baseband signal to quadrature detection and outputs in-phase component (I-component) data and quadrature-component (Q-component) data. The quadrature detector 12 includes a receive-carrier generator 12a, a phase shifter 12b for shifting the phase of the receive carrier by π/2, and multipliers 12c, 12d for multiplying the baseband signal by the receive carrier and outputting the I-component signal and the Q-component signal. Low-pass filters (LPF) 13a, 13b limit the bands of these output signals and AD converters 15a, 15b convert the I- and Q-component signals to digital signals and input the digital signals to a searcher 16, fingers 17a1 to 17a4 and a reception power measurement unit 18.
When a direct-sequence signal (DS signal) that has been influenced by multipath is input to the searcher 16, the latter performs an autocorrelation operation using a matched filter (not shown), thereby detecting multipath, and inputs despreading-start timing data and delay-time adjustment data of the respective paths to the fingers 17a1 to 17a4 corresponding to the respective paths. A despreader/delay-time adjusting unit 21 of each of the fingers 17a1 to 17a4 subjects a direct wave or a delayed wave that arrives via a prescribed path to despread processing using a code identical with the spreading code, performs dump integration, then applies delay processing conforming to the path and outputs two types of signals, namely a pilot signal (reference signal) and information signal. A phase compensator (channel estimation unit) 22 averages the voltages of the I- and Q-components of the pilot signal over a prescribed number of slots and outputs channel estimation signals It, Qt. A synchronous detector 23 restores the phases of despread information signals I′, Q′ based upon a phase difference θ between a pilot signal contained in the receive signal and an already known pilot signal. That is, since the channel estimation signals It, Qt are cosine and sine components of phase difference θ, the synchronous detector 23 performs demodulation (synchronous detection) of receive information signals (I,Q) by applying phase rotation processing to reception information signals (I′,Q′) in accordance with the following equation using the channel estimation signals                                           (                                          I                t                            ,                              Q                t                                      )                    :                      (                                                            I                                                                              Q                                                      )                          =                              (                                                            It                                                  Qt                                                                                                  -                    Qt                                                                    It                                                      )                    ⁢                      (                                                                                I                    ′                                                                                                                    Q                    ′                                                                        )                                              (        1        )            
A RAKE combiner 17b combines the signals output from the fingers 17a1 to 17a4, a multiplier 17d multiplies the combined output of the RAKE combiner by a weighting that conforms to the reception power and outputs the weighted signal, a maximal ratio combiner 19 combines the outputs of respective branches at a ratio that conforms to the size of reception power, and a discrimination unit 20 performs data discrimination based upon the output of the maximal ratio combiner.
With DS-CDMA, all users (all channels) employ the same frequency band in communication with the base station. Consequently, in a case where mobile stations transmit to a base station, a so-called near-far problem occurs. Specifically, if a mobile station near the base station and a mobile station far from the base station transmit at the same power, the transmission power of the nearby mobile station will be more than necessary and will interfere with transmission from the other mobile station. Therefore, in the uplink for mobile-station transmission/base-station reception, the usual practice is to exercise transmission power control for controlling the transmission power of each mobile station in such a manner that reception power will be constant at the base station.
FIG. 25 is a diagram useful in describing uplink-channel closed-loop transmission power control. Here a mobile station 1 includes a spread-spectrum modulator 1a for spread-spectrum modulating transmit data using a spreading code conforming to a prescribed channel specified by a base station, and a power amplifier 1b for amplifying a signal, which is input thereto following processing such as quadrature modulation and frequency conversion applied after spread-spectrum modulation, and transmitting the amplified signal to a base station 2 from an antenna. The base station 2 includes despreaders 2a of respective fingers conforming to the respective paths for applying despread processing to a delay signal that arrives via the assigned path, and a RAKE demodulator 2b for combining the signals output from the fingers, subjecting the combined signal to maximal ratio combining at a weighting conforming to the reception power of each branch, and discriminating “1”s and “0”s of the receive data based upon the maximal-ratio combination signal.
A SIR measurement unit 2c measures the power ratio (SIR: Signal Interference Ratio) of the receive signal (Signal) to an interference signal (Interference), which includes thermal noise. (a) of FIG. 26 shows an example of the SIR measurement unit 2c. A signal-point position altering unit 2c1 which, as shown in (b) of FIG. 26, converts a position vector R (whose I and Q components are RI and RQ, respectively) of a reference (pilot) in the I-jQ complex plane to a point in the first quadrant of the plane. More specifically, the signal-point position altering unit 2c1 takes the absolute values of the I component (in-phase component) RI and Q component (quadrature component) RQ of the position vector R of the received signal point to convert this position vector to a signal in the first quadrant of the I-jQ complex plane. An averaging arithmetic unit 2c2 for calculating the average value m of M symbols of the reference signal included in one slot, a desired wave power arithmetic unit 2c3 for calculating m2 (the power S of the desired signal) by squaring the I and Q components of the average value m and summing the squares, and a reception power calculation unit 2c4 for squaring the I and Q components RI, RQ of the position vector of the reference signal and summing the squares, i.e., for performing the following calculation:P=RI2+RQ2  (2)to thereby calculate the reception power P. An average-value arithmetic unit 2c5 calculates the average value of reception power, and a subtractor 2c6 subtracts m2 (the power S of the desired wave) from the average value of the reception power, thereby outputting interference wave power I. A SIR arithmetic unit 2c7 calculates the SIR from the desired wave power S and interference wave power I in accordance with the equationSIR=S/I  (3)
With reference again to FIG. 25, a comparator 2d compares the measured SIR with a target SIR, creates a command which lowers the transmission power using a TPC (Transmission Power Control) bit if the measured SIR is greater than the target SIR, and creates a command to raise the transmission power using the TPC bit if the measured SIR is less than the target SIR. The target SIR is a SIR value necessary to obtain a BER of, e.g., 10−3 (error occurrence at a rate of one error per 1000). The target SIR is input to the comparator 2d from a target-SIR setting unit 2e. A spread-spectrum modulator 2f spread-spectrum modulates the transmit data and TPC bits. After spread-spectrum modulation, the base station 2 executes processing such as DA conversion, quadrature modulation, frequency conversion and power amplification and transmits the results to the mobile station 1 from an antenna. A despreader 1c in the mobile station 1 applies despread processing to the signal received from the base station 2, and a RAKE demodulator id demodulates the receive data and TPC bits and controls the transmission power of the power amplifier 1b in accordance with a command specified by the TPC bit.
The mobile station 1 and base station 2 perform the above-described transmission power control on a per-slot basis (a) of FIG. 27 is a diagram useful in describing frame/slot structure of an uplink signal from the mobile station 1 to the base station 2. One frame (10 ms) is composed of 16 625-μs slots S0 to S15, each of which consists of, e.g., ten symbols. Each slot constituting a frame for an I component transmits 10 symbols of information, and each slot constituting a frame for a Q component transmits six symbols of a reference signal (pilot) and other signals. The SIR measurement unit 2c measures, slot by slot, the SIR using the six-symbol reference signal contained in each slot of the Q-component frame, and the comparator 2d creates the transmission-power control command using the TCP bit, as mentioned above, in conformity with the comparison between the measured SIR and the target SIR. The base station 2 transmits this transmission-power control command to the mobile station 1 every 625 μs, as shown in (b) of FIG. 27, and the mobile station 1 controls the transmission power in accordance with this command. Since the control cycle is Tslot=0.625 ms, control is capable of following up momentary fluctuation.
When the sending and receiving of voice is considered, it is appropriate to control transmission power upon setting the target SIR so as to obtain a BER on the order of 10−3. If the traveling speed of the mobile station 1 is constant in this case, the BER of 10−3 can be achieved by transmission power control even if the target SIR is fixed. However, if the traveling speed of the mobile station varies and the rate of change in fading increases [i.e., if fading pitch (Hz) rises], then transmission power control based upon the TPC bit can no longer follow up the change in fading. In addition, channel estimation becomes erroneous and BER=10−3 can no longer be maintained.
Further, RAKE gain differs depending also upon the level difference between receive signals on the paths of multiple paths. With conventional transmission power control, a problem which arises is that BER=10−3 can no longer be maintained owing to the number of paths or the level differences between paths. The reason why RAKE gain varies is that when the level of a certain path falls owing to fading, the levels of other paths rise to make up for the fall but this effect depends upon the number of paths and the level differences between paths.